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ISL6269B
Data Sheet August 7, 2006 FN6280.1
High-Performance Notebook PWM Controller with Audio-Frequency Clamp
The ISL6269B IC is a Single-Phase Synchronous-Buck PWM controller featuring Intersil's Robust Ripple Regulator (R3) technology that delivers truly superior dynamic response to input voltage and output load transients. Integrated MOSFET drivers and bootstrap diode result in fewer components and smaller implementation area. Intersil's R3 technology combines the best features of fixedfrequency PWM and hysteretic PWM while eliminating many of their shortcomings. R3 technology employs an innovative modulator that synthesizes an AC ripple voltage signal VR, analogous to the output inductor ripple current. The AC signal VR enters a window comparator where the lower threshold is the error amplifier output VCOMP, and the upper threshold is a programmable voltage reference VW, resulting in generation of the PWM signal. The voltage reference VW sets the steadystate PWM frequency. Both edges of the PWM can be modulated in response to input voltage transients and output load transients, much faster than conventional fixedfrequency PWM controllers. Unlike a conventional hysteretic converter, the ISL6269B has an error amplifier that provides 1% voltage regulation at the FB pin. The ISL6269B has a 1.5ms digital soft-start and can be started into a pre-biased output voltage. A resistor divider is used to program the output voltage setpoint. The ISL6269B can be configured to operate in continuous-conductionmode (CCM) or diode-emulation-mode (DEM), which improves light-load efficiency. In CCM the controller always operates as a synchronous rectifier however, when DEM is enabled the low-side MOSFET is permitted to stay off, blocking negative current flow into the low-side MOSFET from the output inductor.
Features
* High performance R3 technology * Fast transient response * 1% regulation accuracy: -10C to +100C * Wide input voltage range: +7.0V to +25.0V * Output voltage range: +0.6V to +3.3V * Wide output load range: 0A to 25A * Selectable diode emulation mode for increased light load efficiency * Programmable PWM frequency: 200kHz to 600kHz * Pre-biased output start-up capability * Integrated MOSFET drivers and bootstrap diode * Internal digital soft-start * Power good monitor * PWM minimum frequency above audible spectrum * Fault protection - Undervoltage protection - Soft crowbar overvoltage protection - Low-side MOSFET r DS(on) overcurrent protection - Over-temperature protection - Fault identification by PGOOD pull down resistance * Pb-free plus anneal available (RoHS compliant)
Applications
* PCI express graphical processing unit * Auxiliary power rail * VRM * Network adapter
Pinout
ISL6269B (4x4 QFN) TOP VIEW
PGOOD PHASE BOOT
Ordering Information
PART PART NUMBER MARKING TEMP (C) ISL6269BCRZ (See Note)
12 PVCC
UG
PACKAGE
PKG. DWG. #
16 VIN VCC FCCM EN 1 2 3 4 5 COMP
15
14
13
6269BCRZ -10 to +100 16 Ld 4x4 QFN L16.4x4 (Pb-Free) L16.4x4
GND
11 LG 10 PGND 9 ISEN
ISL6269BCRZ-T 6269BCRZ 16 Ld 4x4 QFN Tape and (See Note) Reel (Pb-Free)
6 FB
7 FSET
8 VO
NOTE: Intersil Pb-free plus anneal products employ special Pb-free material sets; molding compounds/die attach materials and 100% matte tin plate termination finish, which are RoHS compliant and compatible with both SnPb and Pb-free soldering operations. Intersil Pb-free products are MSL classified at Pb-free peak reflow temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC J STD-020.
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright Intersil Americas Inc. 2006. All Rights Reserved All other trademarks mentioned are the property of their respective owners.
Block Diagram
VIN VO GND PACKAGE BOTTOM PWM FREQUENCY CONTROL VREF FSET
VCC
EN
gmVIN
VW
+ -
+
PWM
-
OVP
UVP
+
EA FB COMP ISEN IOC DRIVER POR DIGITAL SOFT-START PWM CONTROL PHASE SHOOT THROUGH PROTECTION PVCC UG
-
OCP
30
90
PGOOD
FIGURE 1. SCHEMATIC BLOCK DIAGRAM
-
- +
gmVO
+ -
CR
VCOMP
+
BOOT
60
DRIVER
150OT PGND FCCM
FN6280.1 August 7, 2006
-
+
VR
-
R Q S
+ -
+
+
-
2
ISL6269B
LG
ISL6269B Typical Application
ISL6269B
VIN 7V-25V PGOOD VIN
RPGOOD
CIN
5V PVCC UG
QHIGH_SIDE
RVCC VCC
BOOT
CPVCC
CVCC GND
CBOOT PHASE
LOUT
VOUT 0.6V-3.3V
RSEN FCCM ISEN
COUT
QLOW_SIDE EN LG
RCOMP COMP PGND
CCOMP1 FB CCOMP2 RBOTTOM RTOP VO
FSET RFSET CFSET
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FN6280.1 August 7, 2006
ISL6269B
Absolute Voltage Ratings
ISEN, VIN to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V VCC, PGOOD to GND . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7.0V PVCC to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7.0V GND to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +0.3V EN, FCCM . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to GND, VCC +3.3V VO, FB, COMP, FSET . . . . . . . . . . . . . . . -0.3V to GND, VCC +0.3V PHASE to GND (DC) . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +28V (<100ns Pulse Width, 10J) . . . . . . . . . . . . . . . . . . . . . . . . . -5.0V BOOT to GND, or PGND . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V BOOT to PHASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +7V UG (DC) . . . . . . . . . . . . . . . . . . . . . . .-0.3V to PHASE, BOOT +0.3V (<200ns Pulse Width, 20J) . . . . . . . . . . . . . . . . . . . . . . . . -4.0V LG (DC) . . . . . . . . . . . . . . . . . . . . . . . .-0.3V to PGND, PVCC +0.3V (<100ns Pulse Width, 4J) . . . . . . . . . . . . . . . . . . . . . . . . . . -2.0V
Thermal Information
Thermal Resistance (Typical, Notes 1, 2) JA (C/W) JC (C/W) QFN Package. . . . . . . . . . . . . . . . . . . . 48 11.5 Junction Temperature Range. . . . . . . . . . . . . . . . . .-55C to +150C Operating Temperature Range . . . . . . . . . . . . . . . .-10C to +100C Storage Temperature . . . . . . . . . . . . . . . . . . . . . . . .-65C to +150C Lead Temperature (Soldering, 10s) . . . . . . . . . . . . . . . . . . . . +300C
Recommended Operating Conditions
Ambient Temperature Range. . . . . . . . . . . . . . . . . . . -10C to 100C Supply Voltage (VIN to GND) . . . . . . . . . . . . . . . . . . . . . . 7V to 25V VCC to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5V 5% PVCC to PGND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5V 5%
CAUTION: Stress above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied.
NOTES: 1. JA is measured in free air with the component mounted on a high effective thermal conductivity test board with "direct attach" features. See Tech Brief TB379. 2. For JC, the "case temp" location is the center of the exposed metal pad on the package underside.
Electrical Specifications
PARAMETER VIN VIN Input Bias Current VIN Shutdown Current VCC and PVCC VCC Input Bias Current VCC Shutdown Current PVCC Shutdown Current VCC POR THRESHOLD
These specifications apply for TA = -10C to +100C, unless otherwise stated. All typical specifications TA = +25C, VCC = 5V, PVCC = 5V SYMBOL TEST CONDITIONS MIN TYP MAX UNIT A A A
IVIN
EN = 5V, VIN = 7V EN = 5V, VIN = 25V
-
6.5 26 0.1
10 35 1.0
IVIN_SHDN EN = GND, VIN = 25V
IVCC
EN = 5V, FCCM = GND, FB = 0.65V, VIN = 7V to 25V
-
1.7 0.1 0.1
2.5 1.0 1.0
mA A A
IVCC_SHDN EN = GND, VCC = 5V IPVCC_SHDN EN = GND, PVCC = 5V
Rising VCC POR Threshold Voltage Falling VCC POR Threshold Voltage REGULATION Reference Voltage Regulation Accuracy PWM Frequency Range Frequency-Set Accuracy VO Range VO Input Leakage ERROR AMPLIFIER FB Input Bias Current COMP Source Current COMP Sink Current COMP High Clamp Voltage
VVCC_THR V
VCC_THF
4.35 4.10
4.45 4.20
4.55 4.30
V V
VREF FB connected to COMP
-1
0.6 -
+1
V %
FSW FAUDIO
FCCM = 5V FCCM = GND FSW = 300kHz
200 19 -12 0.60
28 1.3 7.0
600 +12 3.30 -
kHz kHz % V A A A mA mA V
VVO IVO VO = 0.60V VO = 3.30V
-
IFB
FB = 0.60V
-0.5 3.10
2.5 0.3 3.40
+0.5 3.65
ICOMP_SRC FB = 0.40V, COMP = 3.20V ICOMP_SNK FB = 0.80V, COMP = 0.30V VCOMP_HC FB = 0.40V, Sink 50A
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ISL6269B
Electrical Specifications
PARAMETER COMP Low Clamp Voltage POWER GOOD RPG_SS PGOOD Pull Down Impedance RPG_UV RPG_OV RPG_OC PGOOD Leakage Current PGOOD Maximum Sink Current (Note 3) PGOOD Soft-Start Delay GATE DRIVER UG Pull-Up Resistance UG Source Current (Note 3) UG Sink Resistance UG Sink Current (Note 3) LG Pull-Up Resistance LG Source Current (Note 3) LG Sink Resistance LG Sink Current (Note 3) UG to LG Deadtime LG to UG Deadtime BOOTSTRAP DIODE Forward Voltage Reverse Leakage CONTROL INPUTS EN High Threshold EN Low Threshold FCCM High Threshold FCCM Low Threshold EN Leakage VENTHR VENTHF VFCCMTHR VFCCMTHF IENL IENH IFCCML IFCCMH EN = 0V EN = 5.0V FCCM = 0V FCCM = 5.0V 2.0 2.0 0.1 0.1 0.1 2.0 1.0 1.0 1.0 1.0 1.0 V V V V A A A A A A % % % C C VF IR PVCC = 5V, IF = 2mA VR = 25V 0.58 0.2 V A RUGPU IUGSRC RUGPD IUGSNK RLGPU ILGSRC RLGPD ILGSNK tUGFLGR tLGFUGR 200mA Source Current UG - PHASE = 2.5V 250mA Sink Current UG - PHASE = 2.5V 250mA Source Current LG - PGND = 2.5V 250mA Sink Current LG - PGND = 2.5V UG falling to LG rising, no load LG falling to UG rising, no load 1.0 2.0 1.0 2.0 1.0 2.0 0.5 4.0 21 14 1.5 1.5 1.5 0.9 A A A A ns ns TSS EN High to PGOOD High IPGOOD PGOOD = 5mA Sink PGOOD = 5mA Sink PGOOD = 5mA Sink PGOOD = 5mA Sink PGOOD = 5V 75 75 50 25 2.20 95 95 63 32 0.1 5.0 2.75 125 125 85 45 1.0 3.30 A mA ms These specifications apply for TA = -10C to +100C, unless otherwise stated. All typical specifications TA = +25C, VCC = 5V, PVCC = 5V (Continued) SYMBOL TEST CONDITIONS MIN 0.09 TYP 0.15 MAX 0.21 UNIT V
VCOMP_LC FB = 0.80V, Source 50A
FCCM Leakage PROTECTION ISEN OCP Threshold ISEN Short-Circuit Threshold UVP Threshold OVP Rising Threshold OVP Falling Threshold OTP Rising Threshold (Note 3) OTP Hysteresis (Note 3) NOTE: 3. Guaranteed by characterization.
IOC ISC VUV VOVR VOVF TOTR TOTHYS
ISEN sourcing ISEN sourcing
19 81 113 100 -
26 50 84 116 103 150 25
33 87 119 106 -
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FN6280.1 August 7, 2006
ISL6269B Functional Pin Descriptions
GND (Bottom Pad) Signal common of the IC. Unless otherwise stated, signals are referenced to the GND pin, not the PGND pin. VIN (Pin 1) The VIN pin measures the converter input voltage which is a required input to the R3 PWM modulator. Connect across the drain of the high-side MOSFET to the GND pin. VCC (Pin 2) The VCC pin is the input bias voltage for the IC. Connect +5V from the VCC pin to the GND pin. Decouple with at least 1F of a MLCC capacitor from the VCC pin to the GND pin. FCCM (Pin 3) The FCCM pin configures the controller to operate in forcedcontinuous-conduction-mode (FCCM) or diode-emulationmode (DEM.) DEM is disabled when the FCCM pin is pulled above the rising threshold voltage VFCCMTHR, conversely DEM is enabled when the FCCM pin is pulled below the falling threshold voltage VFCCMTHF. EN (Pin 4) The EN pin is the on/off switch of the IC. The soft-start sequence begins when the EN pin is pulled above the rising threshold voltage VENTHR and VCC is above the power-on reset (POR) rising threshold voltage VVCC_THR . When the EN pin is pulled below the falling threshold voltage VENTHF PWM immediately stops. COMP (Pin 5) The COMP pin is the output of the control-loop error amplifier. Compensation components for the control-loop connect across the COMP and FB pins. FB (Pin 6) The FB pin is the inverting input of the control-loop error amplifier. The converter output voltage regulates to 600mV from the FB pin to the GND pin. Program the desired output voltage with a resistor network connected across the VO, FB, and GND pins. Select the resistor values such that FB to GND is 600mV when the converter output voltage is at the programmed regulation value. FSET (Pin 7) The FSET pin programs the PWM switching frequency. Program the desired PWM frequency with a resistor and a capacitor connected across the FSET and GND pins. VO (Pin 8) The VO pin measures the converter output voltage and is used exclusively as an input to the R3 PWM modulator. Connect at the physical location where the best output voltage regulation is desired. ISEN (Pin 9) The ISEN pin programs the threshold of the OCP overcurrent fault protection. Program the desired OCP threshold with a resistor connected across the ISEN and PHASE pins. The OCP threshold is programmed to detect the peak current of the output inductor. The peak current is the sum of the DC and AC components of the inductor current. PGND (Pin 10) The PGND pin conducts the turn-off transient current through the LG gate driver. The PGND pin must be connected to complete the pulldown circuit of the LG gate driver. The PGND pin should be connected to the source of the low-side MOSFET through a low impedance path, preferably in parallel with the trace connecting the LG pin to the gate of the low-side MOSFET. The adaptive shootthrough protection circuit, measures the low-side MOSFET gate-source voltage from the LG pin to the PGND pin. LG (Pin 11) The LG pin is the output of the low-side MOSFET gate driver. Connect to the gate of the low-side MOSFET. PVCC (Pin 12) The PVCC pin is the input voltage bias for the LG low-side MOSFET gate driver. Connect +5V from the PVCC pin to the PGND pin. Decouple with at least 1F of an MLCC capacitor across the PVCC and PGND pins. BOOT (Pin 13) The BOOT pin stores the input voltage for the UG high-side MOSFET gate driver. Connect an MLCC capacitor across the BOOT and PHASE pins. The boot capacitor is charged through an internal boot diode connected from the PVCC pin to the BOOT pin, each time the PHASE pin drops below PVCC minus the voltage dropped across the internal boot diode. UG (Pin 14) The UG pin is the output of the high-side MOSFET gate driver. Connect to the gate of the high-side MOSFET. PHASE (Pin 15) The PHASE pin detects the voltage polarity of the PHASE node and is also the current return path for the UG high-side MOSFET gate driver. Connect the PHASE pin to the node consisting of the high-side MOSFET source, the low-side MOSFET drain, and the output inductor. PGOOD (Pin 16) The PGOOD pin is an open-drain output that indicates when the converter is able to supply regulated voltage. Connect the PGOOD pin to +5V through a pull-up resistor.
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FN6280.1 August 7, 2006
ISL6269B Theory of Operation
Modulator
The ISL6269B is a hybrid of fixed frequency PWM control, and variable frequency hysteretic control. Intersil's R3 technology can simultaneously affect the PWM switching frequency and PWM duty cycle in response to input voltage and output load transients. The term "Ripple" in the name "Robust-Ripple-Regulator" refers to the converter output inductor ripple current, not the converter output ripple voltage. The R3 modulator synthesizes an AC signal VR, which is an ideal representation of the output inductor ripple current. The duty-cycle of VR is the result of charge and discharge current through a ripple capacitor CR. The current through CR is provided by a transconductance amplifier gm that measures the VIN and VO pin voltages. The positive slope of VR can be written as:
V RPOS = ( g m ) * ( V IN - V OUT ) (EQ. 1)
Power-On Reset
The ISL6269B is disabled until the voltage at the VCC pin has increased above the rising power-on reset (POR) VCCR threshold voltage. The controller will become once again disabled when the voltage at the VCC pin decreases below the falling POR VCCF threshold voltage.
EN, Soft-Start, and PGOOD
The ISL6269B uses a digital soft-start circuit to ramp the output voltage of the converter to the programmed regulation setpoint at a predictable slew rate. The slew rate of the soft-start sequence has been selected to limit the inrush current through the output capacitors as they charge to the desired regulation voltage. When the EN pin is pulled above the rising EN threshold voltage VENTHR the PGOOD Soft-Start Delay TSS begins and the output voltage begins to rise. The output voltage enters regulation in approximately 1.5ms and the PGOOD pin goes to high impedance once TSS has elapsed.
The negative slope of VR can be written as:
V RNEG = g m V OUT (EQ. 2)
1.5ms VOUT VCC and PVCC
Where gm is the gain of the transconductance amplifier. A window voltage VW is referenced with respect to the error amplifier output voltage VCOMP, creating an envelope into which the ripple voltage VR is compared. The amplitude of VW is set by a resistor connected across the FSET and GND pins. The VR, VCOMP, and VW signals feed into a window comparator in which VCOMP is the lower threshold voltage and VW is the higher threshold voltage. Figure 2 shows PWM pulses being generated as VR traverses the VW and VCOMP thresholds . The PWM switching frequency is proportional to the slew rates of the positive and negative slopes of VR; the PWM switching frequency is inversely proportional to the voltage between VW and VCOMP.
EN
PGOOD 2.75ms
FIGURE 3. SOFT-START SEQUENCE
Ripple Capacitor Voltage CR
Window Voltage VW
Error Amplifier Voltage VCOMP
The PGOOD pin indicates when the converter is capable of supplying regulated voltage. The PGOOD pin is an undefined impedance if VCC has not reached the rising POR threshold VCCR, or if VCC is below the falling POR threshold VCCF. The ISL6269B features a unique faultidentification capability that can drastically reduce troubleshooting time and effort. The pull-down resistance of the PGOOD pin corresponds to the fault status of the controller. During soft-start or if an undervoltage fault occurs, the PGOOD pulldown resistance is 95, or 30 for an overcurrent fault, or 60 for an overvoltage fault.
PWM
FIGURE 2. MODULATOR WAVEFORMS DURING LOAD TRANSIENT
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FN6280.1 August 7, 2006
ISL6269B
TABLE 1. PGOOD PULL-DOWN RESISTANCE CONDITION VCC Below POR Soft Start or Undervoltage Overvoltage Overcurrent PGOOD RESISTANCE Undefined 95 60 30
Diode Emulation
The ISL6269B normally operates in continuous-conductionmode (CCM), minimizing conduction losses by forcing the low-side MOSFET to operate as a synchronous rectifier. An improvement in light-load efficiency is achieved by allowing the converter to operate in diode-emulation-mode (DEM), where the low-side MOSFET behaves as a smart-diode, forcing the device to block negative inductor current flow. The ISL6269B can be configured to operate in DEM by setting the FCCM pin low. Setting the FCCM pin high will disable DEM. Positive-going inductor current flows from either the source of the high-side MOSFET, or the drain of the low-side MOSFET. Negative-going inductor current usually flows into the drain of the low-side MOSFET. When the low-side MOSFET conducts positive inductor current, the phase voltage will be negative with respect to the GND and PGND pins. Conversely, when the low-side MOSFET conducts negative inductor current, the phase voltage will be positive with respect to the GND and PGND pins. Negative inductor current occurs when the output load current is less than 1/2 the inductor ripple current. Sinking negative inductor through the low-side MOSFET lowers efficiency through unnecessary conduction losses. Efficiency can be further improved with a reduction of unnecessary switching losses by reducing the PWM frequency. It is characteristic of the R3 architecture for the PWM frequency to decrease while in diode emulation. The extent of the frequency reduction is proportional to the reduction of load current. Upon entering DEM, the PWM frequency makes an initial step-reduction because of a 33% step-increase of the window voltage V W. The ISL6269B features an audio filter that clamps the minimum PWM frequency to a level beyond human hearing when the output load current becomes low enough. With FCCM pulled low, the converter will automatically enter DEM after the PHASE pin has detected positive voltage, while the LG gate-driver pin is high, for eight consecutive PWM pulses. The converter will return to CCM on the following cycle after the PHASE pin detects negative voltage, indicating that the body diode of the low-side MOSFET is conducting positive inductor current.
MOSFET Gate-Drive Outputs LG and UG
The ISL6269B has internal gate-drivers for the high-side and low-side N-Channel MOSFETs. The LG gate-driver is optimized for low duty-cycle applications where the low-side MOSFET conduction losses are dominant, requiring a low r DS(on) MOSFET. The LG pulldown resistance is small in order to clamp the gate of the MOSFET below the VGS(th) at turnoff. The current transient through the gate at turnoff can be considerable because the switching charge of a low r DS(on) MOSFET can be large. Adaptive shoot-through protection prevents a gate-driver output from turning on until the opposite gate-driver output has fallen below approximately 1V. The dead-time shown in Figure 4 is extended by the additional period that the falling gate voltage stays above the 1V threshold. The high-side gate-driver output voltage is measured across the UG and PHASE pins while the low-side gate-driver output voltage is measured across the LG and PGND pins. The power for the LG gate-driver is sourced directly from the PVCC pin. The power for the UG gate-driver is sourced from a "boot" capacitor connected across the BOOT and PHASE pins. The boot capacitor is charged from a 5V bias supply through a "boot diode" each time the low-side MOSFET turns on, pulling the PHASE pin low. The ISL6269B has an integrated boot diode connected from the PVCC pin to the BOOT pin.
tLGFUGR
tUGFLGR
50% UG
Overcurrent and Short-Circuit Protection
LG 50%
The overcurrent protection (OCP) and short circuit protection (SCP) setpoint is programmed with resistor RSEN that is connected across the ISEN and PHASE pins. The PHASE pin is connected to the drain terminal of the low-side MOSFET. The SCP setpoint is internally set to twice the OCP setpoint. When an OCP or SCP fault is detected, the PGOOD pin will pulldown to 30 and latch off the converter. The fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF.
FIGURE 4. LG AND UG DEAD-TIME
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FN6280.1 August 7, 2006
ISL6269B
The OCP circuit does not directly detect the DC load current leaving the converter. The OCP circuit detects the peak of positive-flowing output inductor current. The low-side MOSFET drain current ID is assumed to be equal to the positive output inductor current when the high-side MOSFET is off. The inductor current develops a negative voltage across the r DS(on) of the low-side MOSFET that is measured shortly after the LG gate-driver output goes high. The ISEN pin sources the OCP sense current ISEN, through the OCP programming resistor RSEN, forcing the ISEN pin to zero volts with respect to the GND pin. The negative voltage across the PHASE and GND pins is nulled by the voltage dropped across RSEN as ISEN conducts through it. An OCP fault occurs if ISEN rises above the OCP threshold current IOC while attempting to null the negative voltage across the PHASE and GND pins. ISEN must exceed IOC on all the PWM pulses that occur within 20s. If ISEN falls below IOC on a PWM pulse before 20s has elapsed, the timer will be reset. An SCP fault will occur within 10s when ISEN exceeds twice IOC. The relationship between ID and ISEN is written as:
I SEN * R SEN = I D * r DS ( on ) (EQ. 3)
Overvoltage Protection
When an OVP fault is detected, the PGOOD pin will pull down to 60 and latch-off the converter. The OVP fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF. The OVP fault detection circuit triggers after the voltage across the FB and GND pins has increased above the rising overvoltage threshold VOVR. Although the converter has latched-off in response to an OVP fault, the LG gate-driver output will retain the ability to toggle the low-side MOSFET on and off, in response to the output voltage transversing the VOVR and VOVF thresholds.
Undervoltage Protection
When a UVP fault is detected, the PGOOD pin will pull down to 95 and latch-off the converter. The fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF. The UVP fault detection circuit triggers after the voltage across the FB and GND pins has fallen below the undervoltage threshold VUV.
The value of RSEN is then written as:
I PP I + -------- * OC * r SP DS ( on ) FL 2 R SEN = -------------------------------------------------------------------------I OC
Over-Temperature
When the temperature of the ISL6269B increases above the rising threshold temperature TOTR, the IC will enter an OTP state that suspends the PWM , forcing the LG and UG gate-driver outputs low. The status of the PGOOD pin does not change nor does the converter latch-off. The PWM remains suspended until the IC temperature falls below the hysteresis temperature TOTHYS at which time normal PWM operation resumes. The OTP state can be reset if the EN pin is pulled below the falling EN threshold voltage VENTHF or if VCC decays below the falling POR threshold voltage V VCC_THF. All other protection circuits function normally during OTP. It is likely that the IC will detect an UVP fault because in the absence of PWM, the output voltage immediately decays below the undervoltage threshold VUV; the PGOOD pin will pulldown to 95 and latch-off the converter. The UVP fault will remain latched until the EN pin has been pulled below the falling EN threshold voltage VENTHF or if VCC has decayed below the falling POR threshold voltage VVCC_THF.
(EQ. 4)
Where: - RSEN () is the resistor used to program the overcurrent setpoint - ISEN is the current sense current that is sourced from the ISEN pin - IOC is the ISEN threshold current sourced from the ISEN pin that will activate the OCP circuit - IFL is the maximum continuous DC load current - IPP is the inductor peak-to-peak ripple current - OCSP is the desired overcurrent setpoint expressed as a multiplier relative to IFL
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ISL6269B
Programming the Output Voltage
When the converter is in regulation there will be 600mV from the FB pin to the GND pin. Connect a two-resistor voltage divider across the VO pin and the GND pin with the output node connected to the FB pin. Scale the voltage-divider network such that the FB pin is 600mV with respect to the GND pin when the converter is regulating at the desired output voltage. The output voltage can be programmed from 600mV to 3.3V. Programming the output voltage is written as:
R BOTTOM V REF = V OUT * -------------------------------------------------R TOP + R BOTTOM (EQ. 5)
REF R2 C1
C2
COMP
R1
EA
FB
+
Where: - VOUT is the desired output voltage of the converter - VREF is the voltage that the converter regulates to between the FB pin and the GND pin - RTOP is the voltage-programming resistor that connects from the FB pin to the VO pin. In addition to setting the output voltage, this resistor is part of the loop compensation network - RBOTTOM is the voltage-programming resistor that connects from the FB pin to the GND pin Beginning with RTOP between 1k to 5k, calculating RBOTTOM is written as:
V REF * R TOP R BOTTOM = -----------------------------------V OUT - V REF (EQ. 6)
QHIGH_SIDE UG PHASE LOUT Gate Drivers QLOW_SIDE LG GND CESR COUT DCR FSET
RFSET R3 Modulator
CFSET
VO VOUT VIN
VIN
Programming the PWM Switching Frequency
The ISL6269B does not use a clock signal to produce PWM. The PWM switching frequency FSW is programmed by the resistor RFSET that is connected from the FSET pin to the GND pin. The approximate PWM switching frequency is written as:
1 F SW = -------------------------K R FSET (EQ. 7)
Estimating the value of RFSET is written as:
1 R FSET = ------------------K * F SW (EQ. 8)
ISL6269B
Where: - FSW is the PWM switching frequency - RFSET is the FSW programming resistor - K = 75 x 10-12 It is recommended that whenever the control loop compensation network is modified, FSW should be checked for the correct frequency and if necessary, adjust RFSET.
FIGURE 5. COMPENSATION REFERENCE CIRCUIT
Compensation Design
The LC output filter has a double pole at its resonant frequency that causes the phase to abruptly roll downward. The R3 modulator used in the ISL6269B makes the LC output filter resemble a first order system in which the closed loop stability can be achieved with a Type II compensation network.
Your local Intersil representative can provide a PC-based tool that can be used to calculate compensation network component values and help simulate the loop frequency response. The compensation network consists of the internal error amplifier of the ISL6269B and the external components R1, R2, C1, and C2 as well as the frequency setting components RFSET, and CFSET, are identified in the schematic Figure 5.
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FN6280.1 August 7, 2006
ISL6269B General Application Design Guide
This design guide is intended to provide a high-level explanation of the steps necessary to create a single-phase power converter. It is assumed that the reader is familiar with many of the basic skills and techniques referenced below. In addition to this guide, Intersil provides complete reference designs that include schematics, bills of materials, and example board layouts. shared by a sufficient quantity of paralleled capacitors so that they operate below the maximum rated RMS current at FSW. Take into account that the rated value of a capacitor can fade as much as 50% as the DC voltage across it increases.
Selection of the Input Capacitor
The important parameters for the bulk input capacitance are the voltage rating and the RMS current rating. For reliable operation, select bulk capacitors with voltage and current ratings above the maximum input voltage and capable of supplying the RMS current required by the switching circuit. Their voltage rating should be at least 1.25 times greater than the maximum input voltage, while a voltage rating of 1.5 times is a preferred rating. Figure 6 is a graph of the input RMS ripple current, normalized relative to output load current, as a function of duty cycle that is adjusted for converter efficiency. The ripple current calculation is written as:
2 2D 2 ( I MAX ( D - D ) ) + x I MAX ----- 12 I IN_RMS = ---------------------------------------------------------------------------------------------------I MAX
Selecting the LC Output Filter
The duty cycle of an ideal buck converter is a function of the input and the output voltage. This relationship is written as:
V OUT D = --------------V IN V OUT * ( 1 - D ) I PP = ------------------------------------F SW * L OUT (EQ. 9)
The output inductor peak-to-peak ripple current is written as:
(EQ. 10)
A typical step-down DC/DC converter will have an IPP of 20% to 40% of the maximum DC output load current. The value of IPP is selected based upon several criteria such as MOSFET switching loss, inductor core loss, and the resistive loss of the inductor winding. The DC copper loss of the inductor can be estimated by:
P COPPER = I LOAD
2
*
(EQ. 14)
Where: - IMAX is the maximum continuous ILOAD of the converter - x is a multiplier (0 to 1) corresponding to the inductor peak-to-peak ripple amplitude expressed as a percentage of IMAX (0% to 100%) - D is the duty cycle that is adjusted to take into account the efficiency of the converter which is written as:
V OUT D = ------------------------V IN EFF
DCR
(EQ. 11)
Where ILOAD is the converter output DC current. The copper loss can be significant so attention has to be given to the DCR selection. Another factor to consider when choosing the inductor is its saturation characteristics at elevated temperature. A saturated inductor could cause destruction of circuit components, as well as nuisance OCP faults. A DC/DC buck regulator must have output capacitance COUT into which ripple current IPP can flow. Current IPP develops a corresponding ripple voltage VPP across COUT, which is the sum of the voltage drop across the capacitor ESR and of the voltage change stemming from charge moved in and out of the capacitor. These two voltages are written as:
V ESR = I PP * E SR (EQ. 12)
In addition to the bulk capacitance, some low ESL ceramic capacitance is recommended to decouple between the drain of the high-side MOSFET and the source of the low-side MOSFET.
NORMALIZED INPUT RMS RIPPLE CURRENT
0.6 0.55 0.5 0.45 0.4 0.35 0.3 0.25 0.2 0.15 0.1 0.05 0
and
I PP V C = -------------------------------------8 * C OUT * F
SW
x=1 x = 0.75 x = 0.50 x = 0.25 x=0
(EQ. 13)
If the output of the converter has to support a load with high pulsating current, several capacitors will need to be paralleled to reduce the total ESR until the required VPP is achieved. The inductance of the capacitor can cause a brief voltage dip if the load transient has an extremely high slew rate. Low inductance capacitors constructed with reverse package geometry are available. A capacitor dissipates heat as a function of RMS current and frequency. Be sure that IPP is 11
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
DUTY CYCLE
FIGURE 6. NORMALIZED RMS INPUT CURRENT FOR x = 0.8
FN6280.1 August 7, 2006
ISL6269B
MOSFET Selection and Considerations
Typically, a MOSFET cannot tolerate even brief excursions beyond their maximum drain to source voltage rating. The MOSFETs used in the power stage of the converter should have a maximum VDS rating that exceeds the sum of the upper voltage tolerance of the input power source and the voltage spike that occurs when the MOSFET switches off. There are several power MOSFETs readily available that are optimized for DC/DC converter applications. The preferred high-side MOSFET emphasizes low switch charge so that the device spends the least amount of time dissipating power in the linear region. Unlike the low-side MOSFET which has the drain-source voltage clamped by its body diode during turn off, the high-side MOSFET turns off with VIN - VOUT - VLacross it. The preferred low-side MOSFET emphasizes low r DS(on) when fully saturated to minimize conduction loss. For the low-side MOSFET, (LS), the power loss can be assumed to be conductive only and is written as:
P CON_LS I LOAD r DS ( on )_LS * ( 1 - D )
2
As an example, suppose the high-side MOSFET has a total gate charge Qg, of 25nC at VGS = 5V, and a VBOOT of 200mV. The calculated bootstrap capacitance is 0.125F; for a comfortable margin select a capacitor that is double the calculated capacitance, in this example 0.22F will suffice. Use an X7R or X5R ceramic capacitor.
Layout Considerations
As a general rule, power should be on the bottom layer of the PCB and weak analog or logic signals are on the top layer of the PCB. The ground-plane layer should be adjacent to the top layer to provide shielding. The ground plane layer should have an island located under the IC, the compensation components, and the FSET components. The island should be connected to the rest of the ground plane layer at one point.
VIAS TO GROUND PLANE GND OUTPUT CAPACITORS SCHOTTKY DIODE LOW-SIDE MOSFETS INPUT CAPACITORS
VOUT INDUCTOR HIGH-SIDE MOSFETS PHASE NODE
(EQ. 15)
For the high-side MOSFET, (HS), its conduction loss is written as:
P CON_HS = I LOAD
2
*
VIN
r DS ( on )_HS * D
(EQ. 16) FIGURE 7. TYPICAL POWER COMPONENT PLACEMENT
For the high-side MOSFET, its switching loss is written as:
V IN * I VALLEY * T ON * F V IN * I PEAK * T OFF * F SW SW P SW_HS = -------------------------------------------------------------------- + ---------------------------------------------------------------2 2 (EQ. 17)
Signal Ground and Power Ground
The bottom of the ISL6269B QFN package is the signal ground (GND) terminal for analog and logic signals of the IC. Connect the GND pad of the ISL6269B to the island of ground plane under the top layer using several vias, for a robust thermal and electrical conduction path. Connect the input capacitors, the output capacitors, and the source of the lower MOSFETs to the power ground plane. PGND (Pin 10) This is the return path for the pull-down of the LG low-side MOSFET gate driver. Ideally, PGND should be connected to the source of the low-side MOSFET with a low-resistance, low-inductance path . VIN (Pin 1) The VIN pin should be connected close to the drain of the high-side MOSFET, using a low resistance and low inductance path. VCC (Pin 2) For best performance, place the decoupling capacitor very close to the VCC and GND pins. PVCC (Pin 12) For best performance, place the decoupling capacitor very close to the PVCC and PGND pins, preferably on the same side of the PCB as the ISL6269B IC.
Where: - IVALLEY is the difference of the DC component of the inductor current minus 1/2 of the inductor ripple current - IPEAK is the sum of the DC component of the inductor current plus 1/2 of the inductor ripple current - TON is the time required to drive the device into saturation - TOFF is the time required to drive the device into cut-off
Selecting The Bootstrap Capacitor
The selection of the bootstrap capacitor is written as:
Qg C BOOT = ----------------------V BOOT (EQ. 18)
Where: - Qg is the total gate charge required to turn on the high-side MOSFET - VBOOT, is the maximum allowed voltage decay across the boot capacitor each time the high-side MOSFET is switched on
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FN6280.1 August 7, 2006
ISL6269B
FCCM (Pin 3), EN (Pin 4), and PGOOD (Pin 16) These are logic inputs that are referenced to the GND pin. Treat as a typical logic signal. COMP (Pin 5), FB (Pin 6), and VO (Pin 8) For best results, use an isolated sense line from the output load to the VO pin. The input impedance of the FB pin is high, so place the voltage programming and loop compensation components close to the VO, FB, and GND pins keeping the high impedance trace short. FSET (Pin 7) This pin requires a quiet environment. The resistor RFSET and capacitor CFSET should be placed directly adjacent to this pin. Keep fast moving nodes away from this pin. ISEN (Pin 9) Route the connection to the ISEN pin away from the traces and components connected to the FB pin, COMP pin, and FSET pin. LG (Pin 11) The signal going through this trace is both high dv/dt and high di/dt, with high peak charging and discharging current. Route this trace in parallel with the trace from the PGND pin. These two traces should be short, wide, and away from other traces. There should be no other weak signal traces in proximity with these traces on any layer. BOOT (Pin 13), UG (Pin 14), and PHASE (Pin 15) The signals going through these traces are both high dv/dt and high di/dt, with high peak charging and discharging current. Route the UG and PHASE pins in parallel with short and wide traces. There should be no other weak signal traces in proximity with these traces on any layer.
Copper Size for the Phase Node
The parasitic capacitance and parasitic inductance of the phase node should be kept very low to minimize ringing. It is best to limit the size of the PHASE node copper in strict accordance with the current and thermal management of the application. An MLCC should be connected directly across the drain of the upper MOSFET and the source of the lower MOSFET to suppress the turn-off voltage spike.
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FN6280.1 August 7, 2006
ISL6269B Quad Flat No-Lead Plastic Package (QFN) Micro Lead Frame Plastic Package (MLFP)
L16.4x4
16 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE (COMPLIANT TO JEDEC MO-220-VGGC ISSUE C) MILLIMETERS SYMBOL A A1 A2 A3 b D D1 D2 E E1 E2 e k L L1 N Nd Ne P 0.25 0.50 1.95 1.95 0.23 MIN 0.80 NOMINAL 0.90 0.20 REF 0.28 4.00 BSC 3.75 BSC 2.10 4.00 BSC 3.75 BSC 2.10 0.65 BSC 0.60 16 4 4 0.60 12 0.75 0.15 2.25 2.25 0.35 MAX 1.00 0.05 1.00 NOTES 9 9 5, 8 9 7, 8 9 7, 8 8 10 2 3 3 9 9 Rev. 5 5/04 NOTES: 1. Dimensioning and tolerances conform to ASME Y14.5-1994. 2. N is the number of terminals. 3. Nd and Ne refer to the number of terminals on each D and E. 4. All dimensions are in millimeters. Angles are in degrees. 5. Dimension b applies to the metallized terminal and is measured between 0.15mm and 0.30mm from the terminal tip. 6. The configuration of the pin #1 identifier is optional, but must be located within the zone indicated. The pin #1 identifier may be either a mold or mark feature. 7. Dimensions D2 and E2 are for the exposed pads which provide improved electrical and thermal performance. 8. Nominal dimensions are provided to assist with PCB Land Pattern Design efforts, see Intersil Technical Brief TB389. 9. Features and dimensions A2, A3, D1, E1, P & are present when Anvil singulation method is used and not present for saw singulation. 10. Depending on the method of lead termination at the edge of the package, a maximum 0.15mm pull back (L1) maybe present. L minus L1 to be equal to or greater than 0.3mm.
All Intersil U.S. products are manufactured, assembled and tested utilizing ISO9000 quality systems. Intersil Corporation's quality certifications can be viewed at www.intersil.com/design/quality
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see www.intersil.com 14
FN6280.1 August 7, 2006

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